111104di.pdf

design

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Edited by Bill Schweber

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Digitally control room light intensity

Donal McNamara, Analog Devices, Limerick, Ireland and

Kieran Kelly, Analog Devices, Limerick, Ireland.

and temperature settings for dif-

ferent rooms depending upon their

mood or whether they are working or re-

laxing. The circuit in Figure 1 controls

the intensity of the artificial light in a

room and monitors the temperature of

two zones. The two main circuit blocks

are the PIC16C67 master controller and

the ADT7516 temperature-sensor inter-

face, which includes a four-channel ADC

and a quad voltage-output DAC. Other

components include a photodiode and

an op amp that monitor the ambient

light; a rotary potentiometer that sets the

light intensity; an LED bar array and dis-

play driver, which indicate the light-in-

tensity setting; a light-dimmer-control

circuit; and a 16

which indicates the temperature of the

two zones.

On power-up, the PIC16C67 config-

ures its ports to control the LCD and the

ADT7516. The ADT7516 has a dual in-

terface, comprising I 2 C and SPI, so the

master communicates in SPI mode. The

ADT7516 operates in SPI mode, once the

controller initializes the LCD.

The ADT7516 senses both its internal

temperature and the temperature of a re-

mote thermal diode (Q 1 configured as a

diode), and the PIC16C67 displays these

temperatures on the LCD. One of the

ADT7516’s analog inputs monitors a po-

tentiometer that you adjust to set the re-

quired light intensity. The PIC controller

reads the potentiometer value from the

ADT7516 and outputs a corresponding

Digitally control room light intensity ...... 103

Circuit tests V COM drivers ............................ 104

Use two picogate devices

for bidirectional level-shifting.................... 106

Simple nanosecond-width pulse

generator provides high performance .. 108

Accurately measure resistance with

less-than-perfect components .................. 110

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two-character LCD,

DAC value. The DAC controls an

LM3914 LED-bar-array controller that

shows the potentiometer setting on the

array. If you set the potentiometer half-

LED BAR

ARRAY

V D D

LM3914

Figure 1

V D D

LED1

LED2

LED3

LED4

LED5

LED6

LED7

LED8

LED9

LED10

A microcontroller, a temperature

sensor with an A/D converter, and a

few other components form a smart

room-light-intensity controller.

GND

V D D

V DD

IC 1

ADT7516

V OUTA

V OUTB

V OUTC

V OUTD

INPUT

DIMMER CIRCUITRY

LAMP

GND

MAINS SUPPLY

400V

TR1AC

MOC3020

V D D

D+/A IN1 /A IN1+

2N3908

EXTERNAL

TEMPERATURE

SENSOR

Q 1

PIC16C67

LCD

R 2

13k

D–/A IN2 /A IN2–

PORT B6

PORT D

L DAC /A IN3

INTERRUPT

R 1

DIAL

R 3

10k

D OUT /ADD

SOA/DIN

16TWO-

CHARACTER

LCD

A IN4

PORT B4

PORT B8

PORT B7

15V

PORT B2

PORT B1

PORT B0

V REFIN

SCL/SCLK

IC 2

OP07

D 1

PHOTODIODE

–15V

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NOVEMBER 11, 2004 | EDN 103

M ANY PEOPLE FAVOR different light

design

ideas

way, for example, then half of the LEDs

turn on, indicating that you want an in-

tensity that is half of what the light source

can deliver.

A second DAC output controls a

DIAC-based (X1) light-dimmer circuit.

This dimmer circuit operates like any

other light dimmer, except that the DAC

controls it instead of a potentiometer. A

photodiode monitors the intensity from

the light bulb. An OP07 amplifies its out-

put and feeds it into one of the

ADT7516’s analog inputs. The PIC con-

troller uses the potentiometer and pho-

todiode values, which the ADT7516 dig-

itizes, to maintain equilibrium between

the light intensity and the required light

setting. If the photodiode reading is less

than the potentiometer setting, the con-

troller increases the dimmer DAC value;

it decreases the dimmer DAC value if the

reading is greater.

One of the features of the ADT7516 is

its round-robin mode, in which it con-

stantly monitors all of its measurement

channels. The master need not initialize

any conversions during its operation; all

it has to do is read back from four value

registers and act according to its pro-

gram. This circuit ensures a constant light

intensity within a room, saving power

when daylight takes over as the main light

source. It also extends the lifetime of a

light bulb, thus saving on maintenance

bills in a large office environment. You

can also extend the application to include

control of air conditioning and to mem-

orize heat and light settings that suit in-

dividuals tastes.

Circuit tests V COM drivers

Soufiane Bendaoud, Analog Devices, San Jose, CA

F LAT-PANEL LCD mon-

300

use, such as NTSC, PAL, or

SECAM. Computers, on

the other hand, typically

refresh the screen at a 75-

Hz rate. A single picture el-

ement, or pixel, on an LCD

screen comprises three

subpixels, one each of red,

green, and blue.

Electrically, the subpix-

els behave like capacitors,

storing a certain voltage

until the next voltage ar-

rives. Changing the volt-

ages on the subpixels, one row at a time,

refreshes the screen. These voltages use

V COM as a reference. The absolute value of

the voltage differences, V COM , represents

the brightness of the subpixels. The video

signal undergoes inversion

on a frame-by-frame basis

to ensure that the time av-

erage of the pixel voltages

is zero, thus preventing

screen burnout. The cir-

cuit of Figure 1 tests the

V COM driver by applying a

square wave to a capacitor

array representing the sub-

pixels in the panel. This

circuit simulates the worst-

case condition, in which all

the subpixels switch on or

off simultaneously. A pair

of high-power, low-on-

resistance MOSFETs gen-

erates the square wave.

A nonoverlapping drive

itors offer excellent

image quality and

more compact form factor

than CRTs—hence, their

steadily increasing popu-

larity. Unfortunately, the

complexity of their manu-

facturing process makes

LCD monitors consider-

ably more expensive than

CRTs. The amplifier

that drives V COM , the

voltage on the backplane of

the LCD panel, must be able to drive large

capacitive loads, deliver high peak output

currents, and maintain a constant output

voltage. This Design Idea describes a

simple test to measure the usefulness of

AD8665

V+

10 nF 10 nF

10 nF

0 TO 8V

SQUARE WAVE

Figure 1

In this typical V COM driver, an array of capacitors simulates the

subpixel load.

an amplifier used as a V COM driver. First,

consider some video theory. Flat-panel

television screens differ in the rate at

which the screen refreshes. The refresh

rate for TVs depends on the standard you

R 1

200

V OUT2

IRF9521

C 1

100 pF

74HC00

IRF4905

IRF730

Figure 2

V OUT1

IRF9521

74HC00

MTP3055

IRF730

An array of NAND gates drives power MOSFETs in this V COM test circuit.

104 EDN | NOVEMBER 11, 2004

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design

ideas

scheme ensures that both MOSFETs do

not turn on at the same time. Otherwise,

simultaneous conduction would give rise

to high shoot-through currents. Figure

2 shows the MOSFETs and the nonover-

lapping drive scheme. The drive scheme

uses high-speed NAND gates. An RC

network at the input of the second

NAND gate controls the nonoverlap de-

lay. The first pair of MOSFETs acts as

predrivers to provide the current needed

to drive the power-MOSFET output

stage. Figure 3 shows the nonoverlap-

ping drive to the gates of the output

stage. Figure 4 shows the instantaneous

peak output current of the AD8565 in

Figure 1 in response to a pulse from the

test circuit.

10V

V OUT2

CH 2=100 nA/DIV

SEL>>

10V

V(R36:1)

V OUT1

CH 1=5V/DIV

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

V(R37:1)

TIME (SEC)

TIME (2 SEC/DIV)

Nonoverlapping gate drive prevents large shoot-through

currents in the output stage of Figure 2’s circuit.

Figure 4

The AD8565 exhibits peak currents greater

than 250 mA in response to a test-circuit pulse.

Use two picogate devices

for bidirectional level-shifting

Bob Marshall, Philips Semiconductors

I N NEW MIXED-VOLTAGE systems, it is

V CC LOW

often necessary to level-shift a control

signal from a high level to a low level.

An open-drain device, such as the

74LVC1G07, easily performs this shift.

However, when a bidirectional signal re-

quires level-shifting, it takes a bit more

circuitry, because simply tying two open-

drain devices pins together generates just

a latch function.

The circuit in Figure 1 shows how to

connect the 74LVC2G241 and 74LVC-

2G07 devices together to shift the signal

at A from a high level to a low

voltage at B and to shift a low lev-

el at B to a higher level at A. The DIR sig-

nal controls the direction of the transfer.

When DIR is low, the A side is the input,

and the B side is output. When DIR is

high, B becomes the input, and A be-

comes the output. To have B behave as an

input when the DIR signal is low, redo the

circuit so that Pin 3 of the 74LVC2G241

DIR

I/O A

I/O B

LVC2G241

LVC2G07

Figure 1

V CC HIGH

A two-IC circuit allows signal level-shifting in both directions; signal-flow direction is under circuit

control.

becomes the input to Pin 1 of the

74LVC2G07 and Pin 4 of the 74LVC2G07

becomes the input to Pin 2 of the

74LVC2G241.

The highest voltage V CC should supply

the 74LVC2G241, and the lowest voltage

level supply necessary should supply the

74LVC2G07. For example, to shift a sig-

nal from 3.3 to 1.8V, the 1.8V CC should

supply the 74LVC2G07 device. The size

106 EDN | NOVEMBER 11, 2004

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Figure 3

design

ideas

of the pullup resistor is unimportant,

but, for best speed, it should be as small

as practical to reduce the RC change time

of the output signal of the 74LVC2G07.

The current output of the74LVC07A is 24

mA at 3.3V; at that V CC , the pullup resis-

. It should be

as large as possible to reduce power con-

sumption.

The 74LVC2G07 supply level deter-

mines V OL and V OH at B. At 1.8V, the V OH

would be near V CC , and V OL is 0.45V or

lower when driving a 4-mA load. The

74LVC2G07 and 74LVC2G241 provide a

quick and easy way to obtain a bidirec-

tional level translation and take up little

board space.

Simple nanosecond-width pulse generator

provides high performance

Jim Williams, Linear Technology Corp

pulses in response to an input and trig-

ger, such as for sampling applications,

the predictably programmable short-

time-interval generator has broad uses.

The circuit of Figure 1 , built around a

quad high-speed comparator and a high-

speed gate, has settable 0- to 10-nsec out-

put width with 520-psec, 5V transitions.

Pulse width varies less than 100 psec with

5V supply variations of 65%. The mini-

mum input-trigger width is 30 nsec, and

input-output delay is 18 nsec.

Comparator IC 1 inverts the input

pulse ( Figure 2 , Trace A) and isolates the

A=5V/01V

Comparators IC 2 and IC 3 , arranged as

complementary-output-level detectors,

represent the networks’ delay difference as

edge-timing skew. Trace B is IC 3 ’s fixed-

path output, and Trace C is IC 2 ’s variable

output. Gate G 1 ’s output (Trace D), which

is high during IC 2 -IC 3 positive overlap,

presents the circuit output pulse. Figure 2

shows a 5V, 5-nsec width, measured at

50% amplitude, output pulse with

B=5V/01V

C=5V/01V

D=5V/01V

10nSEC/DIV

. The pulse is clean and has well-

defined transitions. Post-transition aber-

rations, within 8%, derive from G 1 ’s

bond-wire inductance and an imperfect

Figure 2

Pulse-generator wave-

forms, viewed in 400-MHz

real-time bandwidth, include input (Trace A),

IC 3 (Trace B) fixed and IC 2 (Trace C) variable

outputs and output pulse (Trace D). RC net-

works differential delay manifests as IC 2 -IC 3

positive overlap. G 1 extracts this interval and

presents circuit output.

termination. IC 1 ’s output drives fixed

and variable RC networks. Programming

resistor R G primarily determines the net-

works’ charge-time difference and, hence,

delay at a scale factor of approx 80

1V/DIV

/nsec.

INPUT

1/4 LT1721

2nSEC/DIV

510

0.8

/nSEC

1/4 LT1721

The 5-nsec-wide output with

IC 1

Figure 3

COMPARATOR

is clean with well-

defined transitions. Post-transition aberrations

are within 8% and derive from G 1 bond-wire

inductance and an imperfect coaxial probe path.

390

VARIABLE

DELAY

8 pF

IC 2

G 1

74AHCO8

AND

GATE

TO 2.5V

0.1

2.5V

DIFFERENTIAL

DELAY

GENERATOR

OUTPUT

1/4 LT1721

620

IC 3

1V/DIV

FIXED

DELAY

8 pF

Figure 1

200pSEC/DIV

This pulse generator has 0- to 10-nsec width and 520-psec transitions. IC 1 unloads termina-

tion and drives the differential delay network. The IC 2 -IC 3 complementary outputs represent

delay difference as edge timing skew. G 1 , which is high during IC 2 -IC 3 ’s positive overlap,

presents circuit output.

The narrowest amplitude

pulse width is 1 nsec, and

the base width measures 1.7 nsec.

Measurement bandwidth is 3.9 GHz.

108 EDN | NOVEMBER 11, 2004

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tor could be as low as 150

I f you need to produce extremely fast

390

Figure 4

design

ideas

coaxial probing path. Figure 3 shows the

narrowest full amplitude, 5V pulse avail-

able. Width measures 1 nsec at the 50%

amplitude point and 1.7 nsec at the base

in a 3.9-GHz bandwidth. Shorter widths

are available if partial amplitude pulses

are acceptable. Figure 4 shows a 3.3V,

700-psec width (50%) with a 1.25-nsec

base. G 1 ’s rise time limits minimum

achievable pulse width. The partial-am-

plitude pulse, 3.3V high, measures 700

psec with a 1.25-nsec base ( Figure 5 ). Fig-

ure 6 , taken in a 3.9-GHz sampled band-

pass, measures 520-psec rise time. Fall

time is similar. The transition of the probe

edge is well-defined and free of artifacts.

1V/DIV

500 pSEC/DIV

200 pSEC/DIV

The partial-amplitude

pulse, 3.3V high, meas-

ures 700 psec wide with a 1.25-nsec base. The

trace granularity is an artifact of the 3.9-GHz-

sampling-oscilloscope operation.

Figure 5

A transition detail in

the 3.9-GHz bandpass

with rise time of 90 psec shows 520-psec rise

time; fall time is similar. The granularity

derives from sampling-oscilloscope operation.

Figure 6

Accurately measure resistance

with less-than-perfect components

Dave Van Ess, Cypress Semiconductor

strain gauges or thermistors,

you must accurately and in-

expensively measure resistance us-

ing circuitry built with imperfect

components and in which

gain and offset errors can

significantly limit the accuracy of

ohmic measurements. The right

circuit topology makes it possible

to eliminate most error terms while

measuring ohms, leaving the accuracy to

be determined by just a single reference

resistor.

Unlike measuring voltage or current,

measuring a passive attribute, such as re-

V REF

V RESPONSE

BUFFER

ADC

current, you need do no math, so

this method was popular when the

cost of computation was more than

the cost of building an accurate

current source. However, the accu-

racy of the current source directly

limits the accuracy of the reading

and any gain or offset errors from

measuring the response voltage off-

sets the accuracy, as well. Addition-

ally, the range of measurement is limited

to the ADC’s signal range, as the follow-

ing equation shows:

R REF

R T

Figure 1

The resistive-divider topology provides a lower cost alterna-

tive to a current source and a precision resistor for calibration.

sistance, requires a stimulus. One method

of measuring resistance is to force a

known current through a resistor and

measure the voltage across the resistor.

Measuring ohms in this way means that,

with the correct selection of stimulus

V 0

V REF+

V 0

V REF+

. . . .

R REF3

R REF2

R REF1

R REF

V 1

BUFFER

G=0.94

12-BIT ADC

R T

. . . .

R T3

R T2

R T1

V 2

V REF

V 2

Figure 2

Figure 3

Remove most gain and offset errors using two measurements

and a ratio calculation.

Extend the idea to handle multiple sensors and signal paths, using

multiplexing through a single buffer and A/D converter.

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